1. Technical Field
The present invention relates to components that extend the frequency range of a Vector Network Analyzer (VNA). More particularly, the present invention relates to high-frequency components such as non-linear transmission lines or shocklines that enable sampler-based VNAs to operate at high frequencies.
2. Related Art
A. High-Frequency Sampler-Based VNA Receivers in General
Sampler-based VNA receivers make use of equivalent-time sampling to down-convert RF stimulus and response signals to lower intermediate-frequency (IF) signals. In effect, the samplers “time-stretch” coupled versions of RF signal waves incident on and reflected from a device under test (DUT). This sampling approach results in a simplified VNA architecture with reduced cost in comparison with one employing fundamental mixing where the RF-to-IF conversion is made using the fundamental local oscillator (LO) signal as opposed to a harmonic of the LO.
FIG. 1 shows a block diagram illustrating typical components of a sampler-based VNA. The RF signal generator 100 provides an RF signal through switch 102 to two possible paths 104 and 105 along which incident signals a1 and a2 are provided to a DUT 106. The RF signal is also coupled through couplers 108a and 110a as an RF reference signal to respective reference samplers 112a and 114a for down-conversion to IF reference signals IFa1 and IFa2. Signals b1 and b2 that are reflected from or transmitted through the DUT 106 are coupled through couplers 108b and 110b to respective test samplers 112b and 114b in the form of test signals for down-conversion to IF signals IFb1 and IFb2. Analog-to-digital converters (not shown) convert the IFa1, IFa2, IFb1 and IFb2 to digital signals for processing and analysis that are geared at extracting the DUT response.
In each of the samplers 112a, 112b, 114a and 114b, the RF signal is mixed with a harmonic of the LO signal generator 120 to form the IF signals IFa1, IFb1, IFa2 and IFb2. The harmonic generator 122 connects LO signal generator 120 to the samplers 112a, 112b, 114a and 114b and provides harmonics of the fundamental LO signal generator 120, thereby increasing significantly the LO frequency provided to the samplers 112a, 112b, 114a and 114b. 
As a direct result of the nature of the equivalent-time-sampling process, the LO source 120 required for strobing the samplers 112a, 112b, 114a and 114b operates in a lower frequency range than would be required in a fundamental-mixer VNA where the LO is directly connected to the mixers. Equivalent-time sampling, however, is provided at the expense of increased conversion loss.
B. Sampler Circuitry
FIG. 2 shows one implementation of a sampling circuit that has been used extensively in microwave VNAs, sampling oscilloscopes, frequency counters, etc. The sampling circuit of FIG. 2 can be used to form samplers 112a, 112b, 114a and 114b of FIG. 1. The circuit of FIG. 2 was introduced by W. M Grove in “Sampling for Oscilloscopes and Other RF Systems: DC Through X-Band,” IEEE MTT, Vol. 66, No. 1, May 1966. In this sampler circuit, a voltage pulse VLO is generated by signal source 200 (which can be formed by the output of harmonic generator 122 of FIG. 1). The VLO source 200 is provided through source resistance 201 (RS-LO). The signal VLO gates the Schottky diodes 202 and 203 over a brief time interval Tg, known also as the gating time. Over this interval, the Schottky diodes 202 and 203 are driven into conduction and result in charging of the sampling capacitors 204 and 206 having a capacitance labeled Cs. The charge present on the capacitors 204 and 206 results in an output waveform provided at VIF through resistors 220 and 222 with value RF that is related to the polarity and amplitude of the RF input from VRF source 208. The signal from RF source 208 is provided through a resistor divider with resistors 210 and 211 each having a resistance Rs. The voltage pulse provided across the sampling bridge (i.e. series connection of 204, 203, 202, 206) is formed by differentiating the step-like voltage waveform generated from the VLO source 200 by means of a pair of commensurate-length shorted stubs (210,212) and (214,218) located on either side of the sampling bridge. To elaborate, transmission lines (212, 210) and (214, 218) are shorted and used to transform a step voltage from VLO source 200 into a pulse that gates the sampling Schottky diodes 202 and 203. The voltage at VIF is further shaped by a filter formed by capacitor 224 (CH) and resistor 226 (RL) connecting the output VIF to ground.
FIG. 3 shows an equivalent circuit for the components of FIG. 2. The equivalent circuit for a Schottky diode is a series combination of a Schottky diode junction resistance Rj 302, its ohmic resistance Rd 304, and an ideal switch gated at the rate of the VLO source. The equivalent circuit of FIG. 3 thus includes the switched gate 300 driven by the VLO source, along with the junction resistance Rj/2 302 and ohmic resistance Rd/2 304 that are equivalent to the combined diodes 202 and 203. The RF source 208 has an equivalent voltage VRF/2 at 308 connected to an equivalent resistance Rs/2 310 for resistors 210 and 211. The capacitors 204 and 206 have an equivalent capacitance 2Cs 306, while filter resistors 220 and 221 have an equivalent resistance RF/2 320. The output filter keeps the same values Cs 324 and RL 326 as elements 224 and 226 of FIG. 2.
The 3-dB RF bandwidth of the sampler shown having the equivalent circuit shown in FIG. 3 will be inversely proportional to the gating time Tg (that is, f3-dBRF≈0.35/Tg). For a given RF frequency fRF, the LO frequency is then chosen to reduce the harmonic number N and, thus, the conversion loss and noise figure of the sampler.
C. Samplers Using Step Recovery Diodes
Practical implementations of samplers for VNAs have relied traditionally on step-recovery diodes (SRD) connected as VLO source 200 to generate pulses applied to the switches. Commercial SRDs are traditionally limited to LO inputs having frequencies that do not exceed a few hundred MHz. This is due to the fact that the transit time of an SRD limits the frequency of its input. This limitation is a fundamental one in the context of microwave and millimeter-wave VNAs since it requires that a high harmonic number N be used in the down-conversion process, resulting in an increase in the noise figure of the sampler due to image-response conversion. In addition, the use of a high harmonic number increases the number of spurious receiver responses and can reduce the effective dynamic range of a VNA.
Another fundamental limitation in an SRD-based VNA is the RF leakage between channels. Because SRDs are fundamentally governed by avalanche phenomena, a single SRD is typically used for all channels of the VNA's receiver. If a separate SRD were used in each channel, the gating pulses would not be synchronous and the phase relationship between the receiver channels would not be stable. As a result, the distribution scheme shown in FIG. 4 is commonly used with a single SRD in order to keep receiver channels synchronous.
FIG. 4 illustrates a VNA configuration with a single SRD circuit 405 to drive the four separate samplers 412a, 412b, 414a and 414b in a two-channel VNA. The SRD 405 is driven by a VLO source 420 with its resistance RS-LO 421, similar to source 200 and resistance 201 of FIG. 2. The signal from the SRD circuit 405 generates pulses that are distributed from power splitter 407 to the four samplers 412a, 412b, 414a and 414b. Each of the samplers can include circuitry similar to that shown in FIG. 2 that is supplied by the VLO source 200 and resistor 201. The reference sampler 412a and test sampler 412b downconvert signals for a first channel A, similar to samplers 212a and 212b of the VNA circuitry of FIG. 2. The reference sampler 414a and test sampler 414b downconvert signals for a second channel B, similar to samplers 214a and 214b of FIG. 2. In channel A, an RF signal is provided from VRF-A source 400a through couplers 408a and 408b to a first port of a highly reflective DUT 406. In channel B, an RF signal from VRF-B source 400a at the second port of DUT 406 is provided through couplers 410a and 410b to a load 400b. The couplers 408a and 408b of channel A provide a similar function to couplers 108a and 108b of FIG. 1. Similarly, couplers 410a and 410b of channel B provide a similar function to couplers 110a and 110b of FIG. 1.
The leakage between channels A and B even with a single SRD 405 can occur in the path between the channels illustrated by dashed lines in FIG. 4. Leakage can occur between samplers 412a, 412b, 414a and 414b, since the SRD output frequency must be high and isolation amplifiers cannot be used. Thus with a purely passive network, there is an isolation limitation (signals leak from one sampler, through the distribution network, into another sampler). Thus, it is desirable to provide other alternatives to create LO signals to drive samplers other than the SRD approach for a high-frequency VNA/measuring transceiver.
D. Samplers Based on Nonlinear Transmission Lines
A nonlinear transmission line (NLTL) provides a distributed alternative to the SRD, thereby providing a VLO signal source 200 for VNA samplers that can operate over a broad range to very high frequencies and experience minimal channel leakage. SRDs made possible the extension of the RF bandwidth in VNAs to 65 GHz. An example of an SRD-based sampling VNA operating to 65 GHz is the Lightning VNA 37397D manufactured by Anritsu Company of Morgan Hill, Calif. But achieving frequencies above 65 GHz using SRDs has been prevented by the limited fall time for the SRD-based samplers. This frequency limitation, however, can be far removed using NLTLs or shocklines.
FIG. 5 shows representative circuit of a sampler-based VNA using NLTLs 561-564 to provide the LO input to samplers 512a, 512b, 514a and 514b. NLTLs are distributed devices that support the propagation of nonlinear electrical waves such as shocks and solitons. As shown by NLTL 561 of FIG. 5, the NLTL is made up of high-impedance transmission line (571,572) loaded periodically with varactor diodes 573 forming a propagation medium whose phase velocity, and thus time delay, is a function of the instantaneous voltage. For a step-like waveform, the trough of the wave travels at a faster phase velocity than the peak, resulting in compression of the fall time, and thus the formation of a steep wave front that approaches that of a shock wave.
Shockline-based samplers, whether used in a VNA or other receivers to achieve very high frequency operation, have been the subject of patents and numerous articles. For example, shockline devices for use in samplers are described in the following: U.S. Pat. No. 5,014,018 entitled “Nonlinear Transmission Line for Generation of Picosecond Electrical Transients,” by Rodwell, et al.; U.S. Pat. No. 7,088,111 entitled “Enhanced Isolation Level Between Sampling Channels in A Vector Network Analyzer,” by K. Noujeim; and U.S. Pat. No. 6,894,581 entitled “Monolithic Nonlinear Transmission Lines and Sampling Circuits with Reduced Shock-Wave-to-Surface-Wave Coupling,” by K. Noujeim.
In contrast with an SRD where output frequencies are limited to tens of GHz, an NLTL can be designed to generate output frequencies spanning hundreds of GHz, making it ideal for gating samplers whose bandwidth exceeds by far that of the aforementioned 65 GHz SRD-based sampler. In fact, it is the NLTL's frequency-scalable input and output that set it apart from SRDs' and allow broadband sampler operation based on lower harmonic numbers, thus resulting in improved noise figure and spurious responses. The input and output frequency ranges of an NLTL are predicted by its input and output Bragg cutoff frequencies, which are a function of the spacing d (shown in NLTL 551) between cells in a shockline as indicated in U.S. Pat. No. 5,014,018 referenced previously. When driven with a sinusoidal signal, such as the VLO signal 520 in FIG. 5, the NLTL circuit compresses the signal's fall time, resulting in a waveform that is rich in high-frequency harmonics. Monolithic implementations of this circuit and derivatives thereof have recently been made on GaAs substrates. See for example, U.S. Pat. No. 4,956,568 entitled “Monolithic Sampler,” by Sy et al; and U.S. Pat. Nos. 5,267,020 and 5,378,939 entitled “Gallium Arsenide Monolithically Integrated Sampling Head Using Equivalent Time Sampling Having a Bandwidth Greater Than 100 GHz,” by Marsland et al. These shockline implementations dealt with the generation of picosecond pulses for the purpose of gating samplers, making possible the down-conversion of extremely high frequency millimeter-wave and submillimeter-wave signals based on the use of lower harmonic numbers, and resulting in the concomitant improvement in noise figure and spurious responses.
FIG. 5 further illustrates that with NLTLs, as opposed to SRDs, a separate one of the NLTLs 561-564 can be used to supply each sampler 512a, 512b, 514a and 514b. Use of separate NLTLs with each sampler does not impact the gating-pulse synchronicity between samplers. This results from the stability of the distributed fall-time compression mechanism in a shockline (or NLTL), and is in sharp contrast with an SRD in which fall time is based on device-dependent charge storage.
FIG. 5 further illustrates that with NLTLs, as opposed to SRDs, isolators 531-534, amplifiers 541-544 and filters 551-554 can be used to improve channel-to-channel isolation. This is possible since amplifiers covering the input frequency range of a shockline provided by the source VLO 520 are feasible. This is in direct contrast with an SRD whose output frequencies are over a range that exceeds that of available amplifiers and isolators. The availability of isolators 531-534, amplifiers 541-544 and filters 551-554 for use with NLTLs is described in U.S. Pat. No. 7,088,111, referenced previously.
Similar to FIG. 4, the circuitry of channel A in FIG. 5 includes an RF signal provided from VRF-A source 500a through couplers 508a and 508b to a first port of DUT 506. In channel B, an RF signal from the VRF-A source 500a at the second port of DUT 506 is provided through couplers 510a and 510b to a load 500b. The couplers 508a and 508b of channel A provide a similar function to couplers 108a and 108b of FIG. 1, and 408a and 408b of FIG. 4. Similarly, couplers 410a and 410b of channel B provide a similar function to couplers 110a and 110b of FIG. 1, and 408a and 408b of FIG. 4. The leakage between channels A and B, though smaller than leakage experienced with the SRD circuitry of FIG. 4, can occur in the path between channels A and B that is illustrated by dashed lines in FIG. 5.
It would be desirable to provide circuitry to make NLTLs or shocklines even more amenable for use in VNAs.